System and method for adaptively matching the frequency response of multiple channels

ABSTRACT

A method for adaptively matching the frequency response of two channels of a received signal includes the steps of receiving a main RF signal on a main antenna, sampling the main RF signal at a sample rate, delaying the sampled main RF signal by a multiple of a sample period, wherein the sample period is the inverse of the sample rate, receiving at least one other RF signal on at least one other auxiliary antenna, sampling each of the at least one other RF signal at the sample rate, filtering each of the at least one other sampled RF signal utilizing an adaptive finite impulse response (FIR) filter having at least one sub-sample-period time delay, and combining the sampled main RF signal with each of the filtered at least one other sampled RF signals.

FIELD OF THE INVENTION

This invention relates generally to radar systems and more particularly,to adaptive digital signal processing and channel frequency responsematching.

BACKGROUND OF THE INVENTION

Radar systems typically use digital signal processing to cancelinterfering (e.g. jamming) signals from a main channel of a radar arrayby adaptively weighting sampled demodulated outputs of one or moreauxiliary channels and subtracting them from the main channel's output.Deep cancellation of jamming signals, however, requires that thefrequency responses of the main and auxiliary channels be highly matchedacross the instantaneous bandwidth. Fixed amplitude and phase offsets donot limit cancellation. Frequency-dependent channel-to-channelmismatches, however, do limit cancellation. These frequency-dependentmismatches may include amplitude ripple, phase ripple, linear amplitudeslope, and linear phase slope, by way of example only. A single adaptiveweight in each auxiliary channel can provide adequate cancellation atthe center of the instantaneous bandwidth, however, the net cancellationmay be inadequate if the bandwidth and post-calibrationfrequency-dependent mismatches are sufficiently large.

Mismatch across a broad instantaneous bandwidth may be mitigated byImplementing an equalizer at the output of each channel's digitaldemodulator to compensate for passband mismatches that are eitherconstant or that change very slowly over long periods of time as theequipment ages or changes temperature. An equalizer may be a finiteimpulse response (FIR) filter, operating at the in-phase/quadrature(I/Q) sample rate, and operative to modify the frequency response ofeach channel so that all channel responses approximately match a commonreference response shape. However, conventional channel equalizationcannot adequately account for the difference in time delay between thejammer signal in one channel and the jammer signal in every otherchannel because these delays continually differ as the radar rotates.

A conventional solution to this problem is to replace the singleadaptive weight in each auxiliary channel with an adaptive FIR filterthat implements a complex adaptive weight in each tap. Adaptivecomputation of these weights will then automatically match the variouschannel responses to maintain deep cancellation across the instantaneousbandwidth, despite changing jammer angles due to array rotation.Typically, the total duration of the FIR is chosen to be greater thanthe inverse of the shortest expected passband ripple period. The totalnumber of taps is then given by this time duration divided by the I/Qsample period T. Depending on the shortest ripple period expected, alarge number of taps may be required, with each tap requiring anadaptive weight. The large number of taps needed to handle a combinationof slow and fast mismatches increases computational complexity andrequires a large amount of jammer signal data to adequately train theadaptive weights. Alternative techniques are desired.

SUMMARY OF THE INVENTION

An embodiment of the present invention involves a method for adaptivelymatching the frequency response of one signal channel to one or moreother signal channels using a number of sample-period delays and anumber of sub-sample period delays that are less than the sample perioddelays. The method includes the steps of delaying the one signal channelby an integer multiple of the sample period, filtering each other signalchannel utilizing an adaptive filter having at least onesub-sample-period (SSP) time delay, and combining the sampled one signalchannel with each of the filtered other signal channels.

An embodiment of the present invention also involves a system foradaptively matching the frequency response of multiple channels,including a main channel configured to receive a main data stream and tooutput a main I/Q data stream at a given sample rate. At least oneauxiliary channel is configured to receive one or more jamming signalsto be removed from the main I/Q data stream and to output an auxiliaryI/Q data stream at the given sample rate. A sub-sample period adaptivefilter for each auxiliary channel is configured to receive the auxiliaryI/Q data stream and output a matched I/Q data stream, and a cancellationnode is configured to cancel the one or more jamming signals from themain I/Q data stream.

BRIEF DESCRIPTION OF THE FIGURES

Understanding of the present invention will be facilitated byconsideration of the following detailed description of the preferredembodiments of the present invention taken in conjunction with theaccompanying drawings, in which like numerals refer to like parts and inwhich:

FIG. 1 shows a block diagram of an adaptive radar system having one ormore sub-sample-period filters according to an embodiment of the presentinvention;

FIG. 2 shows a block diagram of functional components of a sub-sampleperiod adaptive filter of FIG. 1 where T is the I/Q sample period and τ₁through τ_(N1) are sub-sample-period time delays that are less than Tand the W₁, W₂, . . . , are complex adaptive weights;

FIG. 3 shows a block diagram of functional components of an exemplaryall-pass 7-tap FIR filter configured to delay a received data stream by3T+τ according to an embodiment of the present invention where τ is asub-sample-period time delay and α₁ through α₇ are real weights;

FIG. 4 shows a block diagram of functional components of an exemplaryadaptive weight computation device for the adaptive radar system of FIG.1 having a single auxiliary channel (i.e. N=1) according to anembodiment of the present invention; and

FIG. 5 shows a process flow for a sub-sample-period adaptive FIR filtersuitable for use by the adaptive radar system of FIG. 1 according to anembodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

It is to be understood that the figures and descriptions of the presentinvention have been simplified to illustrate elements that are relevantfor a clear understanding of the present invention, while eliminating,for purposes of clarity, other elements found in typical adaptiveprocessing systems and methods (e.g. radar systems). However, becausesuch elements are well known in the art, and because they do notfacilitate a better understanding of the present invention, a discussionof such elements is not provided herein.

FIG. 1 shows an adaptive radar system 100 according to an embodiment ofthe present invention. Adaptive radar system 100 includes a main antenna102, such as a phased array antenna, configured to transmit and receiveradar signals and to output received radio frequency (RF) echo signalsto a downstream main channel receiver 104. Main channel receiver 104 maybe a typical radar receiver configured to receive RF signals from mainantenna 102, amplify the received signals, and output intermediatefrequency (IF) signals to downstream signal processing componentsillustrated collectively in FIG. 1 as main processing block 108. Mainprocessing block 108 may include one or more typical radar processingcomponents/devices configured to receive IF signals from the mainreceiver, perform analog-to-digital (A/D) conversion, digitaldemodulation (DDM), and equalization (i.e. channel matching), and tooutput I/Q samples at an I/Q sample rate of (fs) hertz (Hz) to adownstream main digital beamformer 110. Main digital beamformer 110 maybe a typical digital beamformer configured to receive I/Q samples atf_(S) Hz and to output an I/Q data stream from the main beam or subarraybeam to a downstream noise cancellation node 112. In the exemplaryembodiment, the I/Q sample rate f_(S) Hz may be about 25% greater thanthe bandwidth of the channel (e.g. f_(S)=5 megahertz (MHz) for a radarsystem with a 4 MHz bandwidth).

Adaptive radar system 100 also includes one or more auxiliary channels(1, 2, . . . , N) useful for cancelling out noise or jamming signalsreceived on the main channel. Each auxiliary channel includes anauxiliary antenna 122 _(i) (where i=1, . . . , N) configured to receivetarget free and clutter free signals and to output RF signals to adownstream auxiliary receiver 124 _(i) (where i=1, . . . , N). Auxiliaryantennas 122 _(i) may be separate antennas from main antenna 102 or theymay be subarrays of the main antenna. In an exemplary embodiment, eachsubarray could be the entire antenna aperture with the only differencebetween the main and auxiliary channels being the beam shaping andsteering weights applied to the array elements (i.e. one set of weightsis applied to form the main beam and another set is applied in parallelto form an auxiliary beam). In an alternative exemplary embodiment, eachsubarray may comprise only a single array element. Each auxiliaryreceiver 124 _(i) may be a typical radar receiver configured to receiveRF signals from an auxiliary antenna 122 _(i), amplify the receivedsignals, and output IF signals to downstream signal processingcomponents, illustrated collectively in FIG. 1 as auxiliary processingblock 126, (where i=1, . . . , N). Auxiliary processing block 126 _(i)may include one or more typical radar processing components/devicesconfigured to receive IF signals from the main receiver, perform A/Dconversion, DDM, and equalization (i.e. channel matching), and to outputI/Q samples at a sample rate of f_(S) Hz to a downstream auxiliarydigital beamformer 128 _(i) (where i=1, . . . , N). Auxiliary digitalbeamformer 128 _(i) may be a typical digital beamformer configured toreceive I/Q samples at f_(S) Hz and to output an I/Q data streamdownstream to both an adaptive weight computation device 132 and asub-sample-period (SSP) adaptive filter such as finite impulse response(FIR) filter 130 _(i) (where i=1, . . . , N).

Adaptive weight computation device 132 is a processing componentconfigured to receive a subset of the total available I/Q data streamselected to contain minimal or no target returns and clutter signalreturns and to output a plurality of adaptive weights to each of Ndownstream SSP adaptive FIR filters 130 _(i). By way of example,adaptive weight computation device 132 may be configured to receive I/Qdata streams only during relatively short “listening windows” occurringimmediately before each transmit pulse, thereby minimizing oreliminating target returns and clutter signal returns caused bytransmissions from the main channel. The frequency at which adaptiveweight computation device 132 computes adaptive weights depends onaspects of the radar system. For example, in a fixed radar adaptiveweights may be calculated relatively infrequently (e.g. once percoherent integration period (CIP)). Alternatively, in a rotating radar,adaptive weights may require updating during a CIP (e.g. at the pulserepetition frequency (PRF)). The processing performed by adaptive weightcomputation device 132 is discussed below with reference to FIG. 4.

Each SSP adaptive FIR filter 130 is a processing component configured toreceive both an I/Q data stream at a frequency of f_(S) Hz and one ormore adaptive weights. Each SSP adaptive FIR filter adaptively matchesthe frequency response of the I/Q data stream to that of the mainchannel I/Q data stream using a number of sample period delays and anumber of sub-sample period delays (less than the sample period delays),for output to downstream noise cancellation node 112. The processingperformed by SSP adaptive FIR filter 130 _(i) is discussed below withreference to FIG. 2.

Cancellation node 112 may be a typical adder configured to receive amain channel I/Q data stream contaminated with components of jammingsignal noise, and one or more auxiliary channel I/Q data streamsconfigured to output the various jamming signal components of the mainchannel. The auxiliary channel I/Q data streams from the main channelI/Q data stream, and cancellation node 112 outputs to downstreamcomponents a jam-free I/Q data stream 113. Further downstream componentsmay include typical radar components, such as by way of example only acoherent signal processor 114, a detection and data processor 116, atracker 118, and a command and control processor 120.

FIG. 1 shows SSP adaptive FIR filters 130 _(i) implemented in a digitalbeamforming radar. It should be clear to those of skill in the art,however, that the SSP adaptive FIR filters 130 _(i) may be similarlyimplemented in a conventional radar (e.g. a radar having main digitalbeamformer 110 upstream in the I/Q sample flow from main processingblock 108). Additionally, while the adaptive radar system 100 of FIG. 1performs equalization in the main processing block 108 prior to digitalbeamforming in main digital beamformer 110 _(i) alternative embodimentsof the present invention may omit equalization prior to digitalbeamforming and include a more general SSP adaptive FIR filter 130 _(i)in each auxiliary channel. In such an embodiment, SSP adaptive FIRfilter 130 _(i) may comprise at least several sample-periods and SSPadaptive weights.

FIG. 2 shows an SSP adaptive FIR filter 130 _(i) discussed above withreference to FIG. 1. SSP adaptive FIR filter 130 _(i) is configured toreceive at input 1302 I/Q samples at f_(S) Hz from a correspondingauxiliary digital beamformer 128 _(i) (shown in FIG. 1). A tapped delayline 1304 delays the I/Q sample stream before the I/Q sample stream ismultiplied by each of complex adaptive weights w_(i) indicated generallyas 1306 (where i=1, 2, 3, . . . , N₁+N₂+1). Complex adaptive weightsw_(i) are trained by adaptive weight computation device 132 (discussedbelow with reference to FIG. 4). The products of multiplying the I/Qsample stream by each w_(i) are then summed together by a summer 1308.Summer 1308 outputs to the downstream noise cancellation node 112(FIG. 1) the I/Q sample streams whose channel passband is matched in thesense that the complex frequency response (i.e., amplitude and phase)across the passband has substantially the same shape as the mainchannel's frequency response over the same passband to within apredetermined tolerance.

The tap delay line may include a zero delay tap, N₁ SSP taps (τ), and N₂sample-period taps (T=1/f_(S)). The SSP taps τ may be significantly lessin duration than T. By way of example, in one embodiment of SSP adaptiveFIR filter 130 _(i), T may be 200 nanoseconds (ns) and accordingly 2Tmay be 400 ns. However, τ₁ may be about one nanosecond (ns) and τ₂ maybe two nanoseconds. Thus, a sub-sample-period τ may be hundreds of timessmaller than sample-period T. An all-pass M-tap f_(S) Hz FIR filter 1310_(i) whose frequency response closely approximates unit gain and linearphase slope across the radar's instantaneous bandwidth may beimplemented for delaying the sampled I/Q data stream by τ. An exemplaryFIR filter 1310 _(i) is discussed with reference to FIG. 3 below.

It should be understood by one of ordinary skill in the art that thedelays τ and T shown in FIG. 2 (and in FIGS. 3 and 4 below) are relativedelays. Each SSP adaptive FIR filter 130 _(i) additionally has a bulkdelay, and all I/Q data streams are delayed by this bulk delay to renderdelays of τ and T delays relative to the main channel.

Various applications of the SSP adaptive FIR filter 130 _(i) may requirea mix of SSP taps and sample-period taps as shown, while otherapplications may require only SSP taps. By way of example only, an SSPadaptive FIR filter 103 _(i) configured to replace fixed channelequalization and support adaptive processing, may need a wide range oftap delays to accommodate both slow and fast variations of the multiplechannel responses. By way of alternative example, an SSP adaptive FIRfilter 103 _(i) may be configured in an adaptive radar system 100 havingfixed equalization to remove most of the frequency-response variationsfrom the multiple channels, leaving only time delay differences due tothe changing aspect angle of the radar antenna relative to a set ofjammers as the radar rotates. Such an SSP adaptive FIR filter 103 _(i)may need only a single SSP tap to adaptively match the changing timedelays.

FIG. 3 shows an exemplary all-pass 7-tap f_(S) Hz FIR filter 1310 _(i)whose frequency response closely approximates unit gain and linear phaseslope across the radar's instantaneous bandwidth. FIR filter 1310 _(i)is configured to receive at an input a sampled I/Q data stream Y. Thesampled I/Q data stream Y is representative of an analog signal y(t)sampled at a given sample period T. The FIR filter receives the sampledI/Q data stream and generates at its output a sampled “version” Y′ ofthe analog signal y (t−3T−τ). This sampled version Y′ has the samesample period T, but the underlying analog signal has been shifted intime by the bulk delay (e.g. 3T) and the targeted timed delay (e.g. τ).In the illustrated embodiment, the system is configured to output thesame sampled I/Q data stream delayed by 3T+τ, with 3T being the bulkdelay and τ being the delay relative to the main channel, where τ is onthe order of hundreds of times smaller than the sample period (τ<<T).

Referring still to FIG. 3, FIR filter 1310 includes seven sample period(T) delays 1330. The delayed sampled I/Q data stream is multiplied byfixed real weights 1332 (details of selection of fixed real weights beα₁−α₇ designated as 1332 is discussed with reference to FIG. 5 below).The results of the multiplication with the fixed real weights 1332 areadded by summer 1334, which outputs a sampled I/Q data stream delayedfrom the input by 3T+τ. By delaying the main channel I/Q data stream by3T (i.e. the bulk delay of the exemplary all-pass 7-tap f_(S) Hz FIRfilter 1310 _(i)), a delay in the auxiliary channel relative to the mainchannel of τ is thereby achieved.

Alternative embodiments of the present invention may implement any M-tapI/Q f_(S) Hz FIR filter 1310 _(i) whose frequency response closelyapproximates unit gain and linear phase slope across the radar'sinstantaneous bandwidth. M must be chosen to be an odd integer. Thus, ifthe phase slope dφ/df across the radar's instantaneous bandwidthl is 2π,the group delay of the filter will be ((M−1)/2)T+τ). The linear phase inthe frequency domain is equivalent to a time delay in the time domain.Thus, the main channel may be delayed by (M−1)/2 multiples of T, therebyachieving a delay in the auxiliary channel relative to the main channelof τ. It is appreciated by one of ordinary skill in the art thatnegative delays may be produced by an M-tap I/Q f_(S) FIR filter 1310_(i) since τ represents a relative time increment.

Referring now to FIG. 4, an exemplary adaptive weight computation device132 is shown for an adaptive radar system of FIG. 1 having a singleauxiliary channel (i.e. N=1). Adaptive weight computation device 132 isconfigured to receive at an input an I/Q data stream from the maindigital beamformer 110 as well as an I/Q data stream from the auxiliarydigital beamformer 128 _(i). The streams from the main digitalbeamformer 110 and from auxiliary digital beamformer 128 _(i) areconfigured to receive target-free and clutter-freeinterference-plus-noise I/Q samples. The input I/Q data stream 410 fromthe main channel may be input into an adaptive weight computationprocessor 1322 without any time delay. The input I/Q data stream 420from the auxiliary channel may be input into a tap delay line 1320 whichmay include a zero-delay tap, SSP taps (τ), and sample-period taps (T).The delay tap τ may be significantly less than period T (e.g. on theorder of hundreds of times smaller than period T). In embodiments ofadaptive weight computation devices 132, tap delay line 1320 may be thesame as tap delay line 1304 discussed with reference to FIG. 2 above.Tap delay line 1320 outputs time delayed I/Q data streams into adaptiveweight computation processor 1322. It should be understood by those ofskill in the art that each I/Q data stream is additionally delayed by abulk delay (greater than T), as discussed with reference to FIG. 2above. This bulk delay is an integer multiple of T and is a consequenceof the generation of the subsample period delays (τ), wherein the mainchannel I/Q data stream is delayed by the bulk delay so as to provide anet or relative delay of (τ), between the channels (relative delay ofthe auxiliary channel I/Q data streams with respect to the main channelI/Q data stream).

Adaptive weight computation processor 1322 may be a conventionalprocessor comprising hardware, software, firmware, or any combinationthereof. Adaptive weight computation processor 1322 may implement one ofnumerous known algorithms selected to train the adaptive weights priorto interference cancellation. Adaptive weight computation is well knownin the field of adaptive processing, the details of which are omittedherein for purposes of brevity. Adaptive weight computation processor1322 outputs one or more adaptive weights w_(i) to SSP adaptive FIRfilter 130 ₁.

In alternative embodiments of the present invention, adaptive weightcomputation device 132 may be configured to receive inputs from maindigital beamformer 110 and from N auxiliary digital beamformers 128_((1-N)). Such an adaptive weight computation device 132 would beconfigured to compute one or more adaptive weights for each respectiveinput from an auxiliary digital beamformer 128 _(i) and output to arespective SSP adaptive FIR filter 130 _(i) the one or more adaptiveweights.

Referring now to FIG. 5, there is shown a process flow for generatingthe fixed weights be α₁-α₇ suitable for use by the adaptive FIR filterin FIG. 2 to create the SSP delays τ₁, τ₂, . . . , τ_(n). Thisarchitecture produces an M-sample all-pass time-delay FIR with realcoefficients and a group delay equal to (M−1)/2 I/Q sample periods plusτ, where τ need not be an integer multiple of the I/Q sample period(i.e. τ may be a SSP). A relative time delay of τ is created byfiltering the I/Q data stream to be delayed through the time delay FIRand delaying all other relevant channels by (M−1)/2 I/Q sample periods(i.e. the bulk delay of (M−1)/2 I/Q samples).

At step 502, an M-sample all-pass prototype FIR filter A is createdwhere M is an odd integer. The prototype FIR filter is an all-passfilter, meaning that the I/Q data stream is not modified, but onlydelayed. The prototype FIR filter A includes (M−1)/2 real weights ofzero, a real weight of one, and (M−1)/2 additional real weights of zero.Accordingly, the prototype FIR filter only functions to delay the I/Qdata stream by (M−1)/2 samples (i.e. the group delay). At step 504, anM-point inverse discrete Fourier transform (IDFT) is taken of the Msamples, thereby creating a complex output from step 504. At step 506, alinear phase is applied across the M samples. The slope of the linearphase is proportional to the desired or target sub-sample time delay τ(the linear phase slope is a function of f_(S) and of the time delay τ,thereby creating a phase with slope −2πτ). At step 508, an M-pointdiscrete Fourier transform (DFT) is taken to re-enter the time domain.At step 510, an M-sample amplitude taper is applied to removediscontinuities toward the ends of the sample streams (e.g. the firstand last samples), thereby smoothing out the time-delay ripple. At theoutput of step 510, all of the samples are complex but have a commonphase which is irrelevant to the design of the FIR filter. At step 512,the common phase is removed (e.g. by taking only the real components orparts), resulting in all of the samples (i.e. coefficients) being real.At step 514, the M samples are normalized (e.g. a common amplitudeadjustment is applied to all of the samples), thereby ensuring that thenoise level at the output is the same as the input noise level. Theresult of the process flow is a filter having a delay of(M−1)/(2f_(S))+τ (i.e. ((M−1)/2)T+τ). Thus, the process flow of FIG. 5provides for design of both a bulk delay filter (i.e. the prototype FIRfilter) and for the sub-sample-period relative delay filter (i.e. thefinal product of the process flow providing a delay equal to the bulkdelay+the subsample period τ).

Additional benefits of implementing an adaptive radar having SSPadaptive FIR filters include significant reduction in latency andincreased numerical stability. Still further, fewer training samples arerequired because the number of training samples needed to achieve agiven level of channel match is proportional to the number of adaptiveweights. Finally, enhanced channel match is obtained because a small setof weight training samples is more likely to be statistically stationarythan a larger set.

In embodiments of the present invention, such as the adaptive radarsystem 100 of FIG. 1, N auxiliary channels may be chosen to meet aparticular set of requirements. For example, multiple auxiliary channelsmay be advantageous when the adaptive radar system 100 may encountermultiple simultaneous barrage noise jammers.

Thus, there is disclosed an embodiment of a method for adaptivelymatching the frequency response of multiple channels of a received radarsignal, comprising: receiving a main RF signal on a main antenna;sampling the main RF signal at a given sample rate; delaying the sampledmain RF signal by a given bulk delay equal to an integer number ofsample periods; receiving at least one other RF signal on at least oneother auxiliary antenna associated with the main antenna; sampling eachof the at least one other RF signals at the given sample rate; filteringeach of the other at least one sampled RF signals utilizing an adaptivefinite impulse response (FIR) filter having at least onesub-sample-period time delay less than the sample period of the filteredsignal; and combining the sampled main RF signal with each of thefiltered other sampled RF signals. The sample period is the inverse ofthe sample rate. The sub-sample-period delay is on the order of hundredsof times smaller than the sample period delay. The sub-sample-period maybe a non-integer multiple of the sample period. Each adaptive FIR filtermay further have at least one sample-period delay. Each of thesub-sample-period time delays is generated by one or more fixed weightFIR filters.

In another embodiment a system for adaptively matching the frequencyresponse of multiple channels, comprises: a main channel configured toreceive a main data stream and to output a mainin-phase/quadrature-phase (I/Q) data stream at a given sample rate; oneor more auxiliary channels, each auxiliary channel configured to receiveone or more jamming signals to be removed from the main I/Q data streamand to output an auxiliary I/Q data stream at the given sample rate; anM-tap adaptive finite impulse response (FIR) filter for each auxiliarychannel configured to receive the auxiliary I/Q data stream and tooutput an I/Q data stream; the FIR filter having a first set of tapsseparated by integer multiples of the I/Q sample rate, and a second setof taps separated by sub-sample-period time delays that are less thanthe I/Q sample period, and wherein the M-tap I/Q sample rate FIRcontains M fixed real weights; and an adder configured to sum the outputdata streams of the main channel and each of the auxiliary channels tocancel the one or more jamming signals from the main I/Q data stream.The matched I/Q data stream has a passband matched to the main channelpassband over the instantaneous bandwidth. The sub-sample-period delayis on the order of hundreds of times smaller than the sample perioddelay.

Furthermore, each of the main and auxiliary channels includes a receiverconfigured to receive RF signals from the corresponding main andauxiliary antennas, amplify the received signals, and outputintermediate frequency (IF) signals. Each of the main and auxiliarychannels further includes a digital beamformer downstream of thecorresponding receiver.

In a different embodiment, an adaptive finite impulse response (FIR)filter for filtering an I/Q data stream having a sample period,comprises: a plurality of sample-period tap delays configured to delaythe I/Q data stream by integer multiples of the sample period of the I/Qdata stream; a plurality of sub-sample-period tap delays configured todelay the I/Q data stream by non-integer multiples of the sample periodof the I/Q data stream, the sub-sample period tap delays being at leastan order of magnitude less than the sample period tap delays; a set ofadaptive weights configured to weight samples of the delayed I/Q datastream; and an adder configured to combine the weighted samples of thedelayed I/Q data streams to generate a filtered I/Q data stream. Theweights of the FIR filter are fixed real weights. The sub-sample-periodis a non-integer multiple of the sample period.

In another embodiment, a time delay circuit delays a received I/Q datastream sampled at a sampling rate, comprises a FIR filter comprising anodd integer number of sample period tap delays configured to delay theI/Q data stream, wherein the sample period is the reciprocal of thesampling rate; an odd integer number of fixed real weights configured toweight samples of the delayed I/Q data stream; and an adder configuredto combine the weighted samples of the delayed I/Q data stream. Thegroup delay of the FIR filter is ((M−1)/2)T+τ) where T is the sampleperiod and τ is the sub-sample period delay.

In a further embodiment, a method for determining a set of fixed realweights for a time delay circuit in a computer system including anM-sample all-pass FIR filter, wherein the M samples include M realweights of zero and a center real weight of one, comprises: performingvia a computer processor an M-point inverse discrete Fourier transformof the M samples; applying a liner phase slope corresponding to a timedelay; performing via a computer processor an M-point discrete Fouriertransform of the M samples; applying an M-sample amplitude taper to theM samples; and removing a common phase offset from the M samples.

While the embodiments of the present invention discussed hereingenerally refer to adaptive radar systems, the present invention can beapplied generally to other applications of adaptive processing requiringchannel matching. By way of non-limiting example, the present inventioncould beneficially be implemented in wireless communications systems,including systems using code division multiple access (CDMA) technologyand frequency hopping.

It will be apparent to those skilled in the art that modifications andvariations may be made in the method and system of the present inventionwithout departing from the spirit or scope of the invention. It isintended that the present invention cover the modification andvariations of this invention provided they come within the scope of theappended claims and their equivalents.

What is claimed is:
 1. A method for adaptively matching the frequencyresponse of multiple channels of a received radar signal, comprising:receiving a main RF signal on a main antenna; sampling the main RFsignal at a given sample rate; delaying the sampled main RF signal by agiven bulk delay equal to an integer number of sample periods; receivingat least one other RF signal on at least one other auxiliary antennaassociated with the main antenna; sampling each of the at least oneother RF signals at the given sample rate; filtering each of the otherat least one sampled RF signals utilizing an adaptive finite impulseresponse (FIR) filter having at least one sub-sample-period time delayless than the sample period of the filtered signal; and combining thesampled main RF signal with each of the filtered other sampled RFsignals.
 2. The method of claim 1, wherein the sample period is theinverse of the sample rate.
 3. The method of claim 2, wherein thesub-sample-period delay is on the order of hundreds of times smallerthan the sample period delay.
 4. The method of claim 3, wherein thesub-sample-period is a non-integer multiple of the sample period.
 5. Themethod of claim 4, wherein each adaptive FIR filter further has at leastone sample-period delay.
 6. The method of claim 4, wherein each of thesub-sample-period time delays is generated by one or more fixed weightFIR filters.
 7. A system for adaptively matching the frequency responseof multiple channels, comprising: a main channel configured to receive amain data stream and to output a main in-phase/quadrature-phase (I/Q)data stream at a given sample rate; one or more auxiliary channels, eachauxiliary channel configured to receive one or more jamming signals tobe removed from the main I/Q data stream and to output an auxiliary I/Qdata stream at the given sample rate; an M-tap adaptive finite impulseresponse (FIR) filter for each auxiliary channel configured to receivethe auxiliary I/Q data stream and to output an I/Q data stream; the FIRfilter having a first set of taps separated by integer multiples of theI/Q sample rate, and a second set of taps separated by sub-sample-periodtime delays that are less than the I/Q sample period, and wherein theM-tap I/Q sample rate FIR contains M fixed real weights; and an adderconfigured to sum the output data streams of the main channel and eachof the auxiliary channels to cancel the one or more jamming signals fromthe main I/Q data stream.
 8. The system of claim 7, wherein the matchedI/Q data stream has a passband matched to the main channel passband overthe instantaneous bandwidth.
 9. The system of claim 7, wherein thesub-sample-period delay is on the order of hundreds of times smallerthan the sample period delay.
 10. The system of claim 7, wherein each ofthe main and auxiliary channels includes a receiver configured toreceive RF signals from the corresponding main and auxiliary antennas,amplify the received signals, and output intermediate frequency (IF)signals.
 11. The system of claim 10, wherein each of the main andauxiliary channels further includes a digital beamformer downstream ofsaid corresponding receiver.